FIG. 14 is a main circuit configuration diagram of a heretofore known DC-DC converter, and is described in PTL 1 (identified further on).
In FIG. 14, Ed is a direct current power supply, Q1 to Q4 are MOSFETs (Metal Oxide Semiconductor Field Effect Transistors) acting as semiconductor switching elements, Tr is a transformer, Np is a primary coil of the transformer Tr (the turn number is also assumed to be NO, Np in the same way, is a secondary coil (the turn number is also assumed to be Ns), D1 to D4 are diodes, Sn1 to Sn4 are snubber circuits, Lo is an inductor, and Co is a smoothing capacitor. Also, Vout and Rtn indicate output terminals, Vin a direct current input voltage, and Vo a direct current output voltage.
In FIG. 14, an alternating current voltage generated in the secondary coil Ns of the transformer Tr by switching of the MOSFETs Q1 to Q4 is full-wave rectified by a bridge rectifier circuit formed of the diodes D1 to D4, and thus converted into a direct current voltage. The direct current voltage is smoothed by a smoothing circuit formed of the inductor Lo and smoothing capacitor Co, and output from the output terminals Vout and Rtn.
This heretofore known technology includes the snubber circuits Sn1 to Sn4 in order to suppress surge voltage generated when there is reverse recovery of the diodes D1 to D4. However, there is a problem in that the higher the switching frequency, the greater the increase in resistance loss in the snubber circuits Sn1 to Sn4, and conversion efficiency as a DC-DC converter decreases.
Next, FIG. 15 is a main circuit configuration diagram of a heretofore known resonant DC-DC converter, and is described in PTL 2 and PTL 3 (both identified further on).
In FIG. 15, an inductor Lr and capacitor Cr configuring an LC series resonant circuit are connected to the primary coil Np of the transformer Tr, while other elements are given the same reference signs as in FIG. 14.
In the circuit of FIG. 15, an alternating current voltage generated in the secondary coil Ns of the transformer Tr is full-wave rectified by the bridge rectifier circuit formed of the diodes D1 to D4, and thus converted into a direct current voltage. Further, the direct current voltage is smoothed by the smoothing capacitor Co, and output from the direct current output terminals Vout and Rtn.
This heretofore known technology is characterized in that, as the voltage across the diodes D1 to D4 is clamped at the direct current output voltage when there is reverse recovery of the diodes D1 to D4, the snubber circuits Sn1 to Sn4 shown in FIG. 14 are unnecessary, and conversion efficiency higher than that of the circuit of FIG. 14 is obtained.
Frequency modulation control described in PTL 4 (identified further on) is known as one example of a method of controlling the direct current output voltage of the circuit shown in FIG. 15.
FIG. 16 shows the relationship between a normalized frequency F and a normalized voltage conversion rate M in the case of the frequency modulation control described in PTL 4. Herein, the normalized frequency F is the ratio of a switching frequency Fs of the switching elements Q1 to Q4 of FIG. 15 to a series resonant frequency Fr of the inductor Lr and capacitor Cr, and is expressed by F=Fs/Fr.
Also, the normalized voltage conversion rate M is the ratio of the direct current output voltage Vo to the direct current input voltage Vin (Vo/Vin) multiplied by a turn ratio n=Np/Ns, and is expressed by M=n·Vo/Vin.
The resonant DC-DC converter shown in FIG. 15 is such that the characteristics of the normalized frequency F and normalized voltage conversion rate M change in accordance with the weight of the load, as shown in FIG. 16. In the case of a light load, the normalized voltage conversion rate M does not drop to or below a certain value, regardless of how far the normalized frequency F is increased, because of which the output voltage range is narrow. Consequently, when the resonant DC-DC converter is used in a battery charger or the like, it is difficult to charge a battery that is in an over-discharged state. Phase modulation control described in PTL 2, and a control method whereby there is switching between frequency modulation control and phase modulation control described in PTL 3, are known as ways of resolving the heretofore described problem of the output voltage range being narrow.
FIG. 17 shows the relationship between the normalized frequency F and normalized voltage conversion rate M in the case of the phase modulation control based on PTL 2.
The technology disclosed in PTL 2 is such that, as phase modulation control (phase shift control) is executed with the normalized frequency F as 1, that is, with the switching frequency Fs equivalent to the series resonant frequency Fr, as shown in FIG. 17, the output voltage range of the DC-DC converter is wider than in FIG. 16.
Also, FIG. 18 shows the relationship between the normalized frequency F and normalized voltage conversion rate M in the case of the frequency modulation control and phase modulation control disclosed in PTL 3.
The technology disclosed in PTL 3 is such that, as shown in FIG. 18, frequency modulation control is executed in a range from the normalized frequency F to a maximum frequency Fmax, and with regard to a voltage range in which output is not possible with frequency modulation control, the output voltage range is expanded beyond that in FIG. 16 by switching to phase modulation control whereby the switching frequency Fs is fixed at the maximum frequency Fmax.
Herein, FIG. 19 is a timing chart showing an operation when executing phase modulation control with the circuit shown in FIG. 15 as a target, and is described in PTL 2. The operation is such that, for example, by repeating an operation whereby the MOSFETs Q1 and Q3 are put into an on-state for a period of times t2 to t3 within one cycle T, and the MOSFETs Q2 and Q4 are put into an on-state for a period of times t4 to t5, a period tcom (a commutation period), for which an output voltage Vuv of the full-bridge circuit formed of the MOSFETs Q1 to Q4 is zero, and a period ton (a conduction period), for which the output voltage Vuv is +Vin or −Yin, are generated.
The conduction period ton is a period for which the voltage of the direct current power supply Ed is applied to the series resonant circuit, while the commutation period tcom is a period for which the voltage of the direct current power supply Ed is not applied to the series resonant circuit, and by controlling the conduction period ton by shifting the phases in which the MOSFETs Q1 to Q4 are turned on or off, it is possible to control the direct current output voltage Vo to a predetermined value.